Method for conditioning a digital speech signal

ABSTRACT

The invention concerns a method for conditioning a digital speech signal(s) processed by successive frames, which consists carrying out a harmonic analysis to estimate the pitch on each frame where it has a speech activity, and in oversampling at an oversampling frequency (f e ) which is a multiple of the estimated pitch.

BACKGROUND OF THE INVENTION

The present invention concerns digital speech signal processing techniques.

Many representations of speech signals take account of the harmonic content of such signals resulting from the manner in which they are produced. In most cases, this is reflected in the determination of a pitch frequency of the speech signal.

Digital processing of speech signals has recently expanded greatly in varied domains: speech coding for transmission and storage, speech recognition, noise reduction, echo cancellation, etc. Such processing very frequently uses an estimate of the pitch frequency and particular operations related to the estimated frequency.

Many methods have been developed for estimating the pitch frequency. One method that is routinely used is based on linear prediction which evaluates a prediction delay which is inversely proportional to the pitch frequency. The delay can be expressed as an integer or fractional number of digital signal sample times. Other methods detect directly breaks in the signal which can be attributed to glottal closures of the speaker, the time intervals between such breaks being inversely proportional to the pitch frequency.

If the digital speech signal is transformed into the frequency domain, as by a discrete Fourier transform, it is necessary to consider a discrete spectrum of the speech signal. The discrete frequencies considered are of the form (a/N)×F_(e), where F_(e) is the sampling frequency, N is the number of samples of the blocks used in the discrete Fourier transform and a is an integer from 0 to N/2−1. These frequencies do not necessarily include the estimated pitch frequency and/or its harmonics. This causes inaccuracy in operations relating to the estimated pitch, which can cause distortion of the processed signal, affecting its harmonic character.

A principal object of the present invention is to propose a method of conditioning the speech signal which makes it less sensitive to the above drawbacks.

SUMMARY OF THE INVENTION

The invention therefore proposes a method of conditioning a digital speech signal processed by successive frames, wherein harmonic analysis of the speech signal is performed to estimate a pitch frequency of the speech signal over each frame in which it features vocal activity. After estimating the pitch frequency of the speech signal over one frame, the speech signal of the frame is conditioned by oversampling it at an oversampling frequency which is a multiple of the estimated pitch frequency.

In processing the speech signal, this enables the frequencies closest to the estimated pitch to be favoured over other frequencies. The harmonic character of the speech signal is therefore preserved as far as possible. To compute spectral components of the speech signal, the conditioned signal is distributed between blocks of N samples which are transformed into the frequency domain and the ratio between the oversampling frequency and the estimated pitch frequency is chosen as a factor of the number N.

The foregoing technique can be refined by estimating the pitch frequency of the speech signal over a frame in the following manner:

estimating time intervals between two consecutive breaks of the signal which can be attributed to glottal closures of the speaker occurring during the frame, the estimated pitch frequency being inversely proportional to said time intervals;

interpolating the speech signal in said time intervals, so that the conditioned signal resulting from such interpolation has a constant time interval between two consecutive breaks.

This approach artificially constructs a signal frame over which the speech signal features breaks at constant intervals. Any variations of the pitch over the duration of a frame are therefore taken into account.

In a further improvement, after processing each conditioned signal frame, a number of the signal samples supplied by such processing is retained which is equal to an integer multiple of the ratio between the sampling frequency and the estimated pitch frequency. This avoids the distortion problems caused by phase discontinuities between frames, which are generally not totally corrected by conventional overlap-add techniques.

Using the oversampling technique to condition the signal yields a good measurement of the degree of voicing of the speech signal over the frame, based on the entropy of the autocorrelation of the spectral components computed on the basis of the conditioned signal. The greater the disturbance of the spectrum, i.e. the more it is voiced, the lower the entropy values. Conditioning the speech signal accentuates the irregularity of the spectrum and therefore the entropy variations, with the result that the latter constitutes a measurement of good sensitivity.

In the remainder of this description, the conditioning method according to the invention is illustrated in a system for suppressing noise in a speech signal. Clearly the method can find applications in many other types of digital speech processing: coding, recognition, echo cancellation, etc.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a noise suppression system implementing the present invention;

FIGS. 2 and 3 are flowcharts of procedures used by a vocal activity detector of the system shown in FIG. 1;

FIG. 4 is a diagram representing the states of a vocal activity detection automation;

FIG. 5 is a graph showing variations in a degree of vocal activity;

FIG. 6 is a block diagram of a module for overestimating the noise of the system shown in FIG. 1;

FIG. 7 is a graph illustrating the computation of a masking curve;

FIG. 8 is a graph illustrating the use of masking curves in the system shown in FIG. 1;

FIG. 9 is a block diagram of another noise suppression system implementing the present invention;

FIG. 10 is a graph illustrating a harmonic analysis method that can be used in a method according to the invention; and

FIG. 11 shows part of a variant of the block diagram shown in FIG. 9.

DESCRIPTION OF PREFERRED EMBODIMENTS

The noise suppression system shown in FIG. 1 processes a digital speech signal s. A windowing module 10 formats the signal s in the form of successive windows or frames each made up of a number N of digital signal samples. In the usual way, these frames can overlap each other. In the remainder of this description, the frames are considered to be made up of N=256 samples with a sampling frequency F_(e) of 8 kHz, with Hamming weighting in each window and with 50% overlaps between consecutive windows, although this is not limiting on the invention.

The signal frame is transformed into the frequency domain by a module 11 using a conventional fast Fourier transform (FFT) algorithm to compute the modulus of the spectrum of the signal. The module 11 then delivers a set of N=256 frequency components S_(n,f) of the speech signal, where n is the number of the current frame and f is a frequency from the discrete spectrum. Because of the properties of the digital signals in the frequency domain, only the first N/2=128 samples are used.

Instead of using the frequency resolution available downstream of the fast Fourier transform to compute the estimates of the noise contained in the signal s, a lower resolution is used, determined by a number I of frequency bands covering the bandwidth [0, F_(e)/2] of the signal. Each band i (1≦i≦I) extends from a lower frequency f(i−1) to a higher frequency f(i), with f(0)=0 and f(I)=F_(e)/2. The subdivision into frequency bands can be uniform (f(i)−f(I−1)=F_(e)/2I). It can also be non-uniform (for example according to a barks scale). A module 12 computes the respective averages of the spectral components S_(n,f) of the speech signal in bands, for example by means of a uniform weighting such as: $\begin{matrix} {S_{n,i} = {\frac{1}{{f(i)} - {f\left( {i - 1} \right)}}{\sum\limits_{f \in {\lbrack{{f{({i - 1})}},{f{(i)}}}}}\quad S_{n,f}}}} & (1) \end{matrix}$

This averaging reduces fluctuations between bands by averaging the contributions of the noise in the bands, which reduces the variance of the noise estimator. Also, this averaging greatly reduces the complexity of the system.

The averaged spectral components S_(n,i) are sent to a vocal activity detector module 15 and a noise estimator module 16. The two modules 15, 16 operate conjointly in the sense that degrees of vocal activity γ_(n,i) measured for the various bands by the module 15 are used by the module 16 to estimate the long-term energy of the noise in the various bands, whereas the long-term estimates {circumflex over (B)}_(n,i) are used by the module 15 for a priori suppression of noise in the speech signal in the various bands to determine the degrees of vocal activity γ_(n,i).

The operation of the modules 15 and 16 can correspond to the flowcharts shown in FIGS. 2 and 3.

In steps 17 through 20, the module 15 effects a priori suppression of noise in the speech signal in the various bands i for the signal frame n. This a priori noise suppression is effected by a conventional non-linear spectral subtraction scheme based on estimates of the noise obtained in one or more preceding frames. In step 17, using the resolution of the bands I, the module 15 computes the frequency response Hp_(n,i) of the a priori noise suppression filter from the equation: $\begin{matrix} {{Hp}_{n,i} = \frac{S_{n,i} - {\alpha_{{n - {\tau 1}},i}^{\prime} \cdot {\hat{B}}_{{n - {\tau 1}},i}}}{S_{{n - {\tau 2}},i}}} & (2) \end{matrix}$

where τ1 and τ2 are delays expressed as a number of frames (τ1≧1, τ2≧0) , and α′_(n,i) is a noise overestimation coefficient determined as explained later. The delay τ1 can be fixed (for example τ1=1) or variable. The greater the degree of confidence in the detection of vocal activity, the lower the value of τ1.

In steps 18 to 20, the spectral components Êp_(n,i) are computed from:

Êp _(n,i)=max{Hp _(n,i) ·S _(n,i) ,βp _(i) ·{circumflex over (B)} _(n−τ1,i)}  (3)

where βp_(i) is a floor coefficient close to 0, used conventionally to prevent the spectrum of the noise-suppressed signal from taking negative values or excessively low values which would give rise to musical noise.

Steps 17 to 20 therefore essentially consist of subtracting from the spectrum of the signal an estimate of the a priori estimated noise spectrum, over-weighted by the coefficient α′_(n−τ1,i).

In step 21, the module 15 computes the energy of the a priori noise-suppressed signal in the various bands i for frame n: E_(n,i)=Êp_(n,i) ². It also computes a global average E_(n,0) of the energy of the a priori noise-suppressed signal by summing the energies for each band E_(n,i), weighted by the widths of the bands. In the following notation, the index i=0 is used to designate the global band of the signal.

In steps 22 and 23, the module 15 computes, for each band i (0≦i≦I), a magnitude ΔE_(n,i) representing the short-term variation in the energy of the noise-suppressed signal in the band i and a long-term value {overscore (E)}_(n,i) of the energy of the noise-suppressed signal in the band i. The magnitude ΔE_(n,i) can be computed from a simplified equation: ${\Delta \quad E_{n,i}} = {{\frac{E_{{n - 4},i} + E_{{n - 3},i} - E_{{n - 1},i} - E_{n,i}}{10}}.}$

As for the long-term energy {overscore (E)}_(n,i), it can be computed using a forgetting factor B1 such that 0<B1<1, namely {overscore (E)}_(n,i)=B1·{overscore (E)}_(n−1,i)+(1−B1)·E_(n,i).

After computing the energies E_(n,i) of the noise-suppressed signal, its short-term variations ΔE_(n,i) and its long-term values {overscore (E)}_(n,i) in the manner indicated in FIG. 2, the module 15 computes, for each band i (0≦i≦I), a value ρ_(i) representative of the evolution of the energy of the noise-suppressed signal. This computation is effected in steps 25 to 36 in FIG. 3, executed for each band i from i=0 to i=I. The computation uses a long-term noise envelope estimator ba_(i), an internal estimator bi_(i) and a noisy frame counter b_(i).

In step 25, the magnitude ΔE_(n,i) is compared to a threshold ε1. If the threshold ε1 has not been reached, the counter b_(i) is incremented by one unit in step 26. In step 27, the long-term estimator ba_(i) is compared to the smoothed energy value {overscore (E)}_(n,i). If ba_(i)≧{overscore (E)}_(n,i), the estimator ba_(i) is taken as equal to the smoothed value {overscore (E)}_(n,i) in step 28 and the counter b_(i) is reset to zero. The magnitude ρ_(i), which is taken as equal to ba_(i)/{overscore (E)}_(n,i) (step 36), is then equal to 1.

If step 27 shows that ba_(i)<{overscore (E)}_(n,i), the counter b_(i) is compared to a limit value bmax in step 29. If b_(i)>bmax, the signal is considered to be too stationary to support vocal activity. The aforementioned step 28, which amounts to considering that the frame contains only noise, is then executed. If b_(i)≦bmax in step 29, the internal estimator bi_(i) is computed in step 33 from the equation:

bi _(i)=(1−Bm)·{overscore (E)} _(n,i) +Bm·ba _(i)  (4)

In the above equation, Bm represents an update coefficient from 0.90 to 1. Its value differs according to the state of a vocal activity detector automaton (steps 30 to 32). The state δ_(n−1) is that determined during processing of the preceding frame. If the automaton is in a speech detection state (δ_(n−1)=2 in step 30) , the coefficient Bm takes a value Bmp very close to 1 so the noise estimator is very slightly updated in the presence of speech. Otherwise, the coefficient Bm takes a lower value Bms to enable more meaningful updating of the noise estimator in the silence phase. In step 34, the difference ba_(i)-bi_(i) between the long-term estimator and the internal noise estimator is compared with a threshold ε2. If the threshold ε2 has not been reached, the long-term estimator ba_(i) is updated with the value of the internal estimator bi_(i) in step 35. Otherwise, the long-term estimator ba_(i) remains unchanged. This prevents sudden variations due to a speech signal causing the noise estimator to be updated.

After the magnitudes ρ_(i) have been obtained, the module 15 proceeds to the vocal activity decisions of step 37. The module 15 first updates the state of the detection automaton according to the magnitude ρ₀ calculated for all of the band of the signal. The new state δ_(n) of the automaton depends on the preceding state δ_(n−1) and on ρ₀, as shown in FIG. 4.

Four states are possible: δ=0 detects silence, or absence of speech, δ=2 detects the presence of vocal activity and states δ=1 and δ=3 are intermediate rising and falling states. If the automaton is in the silence state (δ_(n−1)=0), it remains there if ρ₀ does not exceed a first threshold SE1, and otherwise goes to the rising state. In the rising state (δ_(n−1)=1), it reverts to the silence state if ρ₀ is smaller than the threshold SE1, goes to the speech state if ρ₀ is greater than a second threshold SE2 greater than the threshold SE1 and it remains in the rising state if SE1≦ρ₀≦SE2. If the automaton is in the speech state (δ_(n−1)=2), it remains there if ρ₀ exceeds a third threshold SE3 lower than the threshold SE2, and enters the falling state otherwise. In the falling state δ_(n−1)=3), the automaton reverts to the speech state if ρ₀ is higher than the threshold SE2, reverts the silence state if ρ₀ is below a fourth threshold SE4 lower than the threshold SE2 and remains in the falling state if SE4≦ρ₀≦SE2.

In step 37, the module 15 also computes the degrees of vocal activity γ_(n,i) in each band i≧1. This degree γ_(n,i) is preferably a non-binary parameter, i.e. the function γ_(n,i)=g(ρ_(i)) is a function varying continuously in the range from 0 to 1 as a function of the values taken by the magnitude ρ_(i). This function has the shape shown in FIG. 5, for example.

The module 16 calculates the estimates of the noise on a band by band basis, and the estimates are used in the noise suppression process, employing successive values of the components S_(n,i) and the degrees of vocal activity γ_(n,i). This corresponds to steps 40 to 42 in FIG. 3. Step 40 determines if the vocal activity detector automaton has just gone from the rising state to the speech state. If so, the last two estimates {circumflex over (B)}_(n−1,i) and {circumflex over (B)}_(n−2,i) previously computed for each band i≧1 are corrected according to the value of the preceding estimate {circumflex over (B)}_(n−3,i). The correction is done to allow for the fact that, in the rise phase (δ=1), the long-term estimates of the energy of the noise in the vocal activity detection process (steps 30 to 33) were computed as if the signal included only noise (Bm=Bms), with the result that they may be subject to error.

In step 42, the module 16 updates the estimates of the noise on a band by band basis using the equations:

{tilde over (B)} _(n,i)=λ_(B·) {circumflex over (B)} _(n−1,i)+(1−λ_(B))·S _(n,i)  (5)

{circumflex over (B)} _(n,i)=γn,i·{circumflex over (B)} _(n−1,i)+(1−γ_(n,i))·{tilde over (B)}_(n,i)  (6)

in which λ_(B) designates a forgetting factor such that 0<λ_(B)<1. Equation (6) shows that the non-binary degree of vocal activity γ_(n,i) is taken into account.

As previously indicated, the long-term estimates of the noise {circumflex over (B)}_(n,i) are overestimated by a module 45 (FIG. 1) before noise suppression by non-linear spectral subtraction. The module 45 computes the overestimation coefficient α′_(n,i) previously referred to, along with an overestimate {circumflex over (B)}′_(n,i) which essentially corresponds to α′_(n,i·{circumflex over (B)}) _(n,i).

FIG. 6 shows the organisation of the overestimation module 45. The overestimate {circumflex over (B)}′_(n,i) is obtained by combining the long-term estimate {circumflex over (B)}n,i and a measurement ΔB_(n,i) ^(max) of the variability of the component of the noise in the band i around its long-term estimate. In the example considered, the combination is essentially a simple sum performed by an adder 46. It could instead be a weighted sum.

The overestimation coefficient α′_(n,i) is equal to the ratio between the sum {circumflex over (B)}_(n,i)+ΔB_(n,i) ^(max) delivered by the adder 46 and the delayed long-term estimate {circumflex over (B)}_(n−τ3,i) (divider 47), with a ceiling limit value α_(max), for example α_(max)=4 (block 48). The delay τ3 is used to correct the value of the overestimation coefficient α′_(n,i), if necessary, in the rising phases (δ=1), before the long-term estimates have been corrected by steps 40 and 41 from FIG. 3 (for example τ3=3).

The overestimate {circumflex over (B)}′_(n,i) is finally taken as equal to α′n,i·{circumflex over (B)}_(n−τ3,i) (multiplier 49).

The measurement ΔB_(n,i) ^(max) of the variability of the noise reflects the variance of the noise estimator. It is obtained as a function of the values of S_(n,i) and of {circumflex over (B)}_(n,i) computed for a certain number of preceding frames over which the speech signal does not feature any vocal activity in band i. It is a function of the differences |S_(n−k,i)−{circumflex over (B)}_(n−k,i)| computed for a number K of silence frames (n−k≦n). In the example shown, this function is simply the maximum (block 50). For each frame n, the degree of vocal activity γ_(n,i) is compared to a threshold (block 51) to decide if the difference |S_(n,i)−{circumflex over (B)}_(n,i)| calculated at 52-53, must be loaded into a queue 54 with K locations organised in first-in/first-out (FIFO) mode, or not. If γ_(n,i) does not exceed the threshold (which can be equal to 0 if the function g() has the form shown in FIG. 5), the FIFO 54 is not loaded; otherwise it is loaded. The maximum value contained in the FIFO 54 is then supplied as the measured variability ΔB_(n,i) ^(max).

The measured variability ΔB_(n,i) ^(max) can instead be obtained as a function of the values S_(n,f) (not S_(n,i)) and {circumflex over (B)}_(n,i). The procedure is then the same, except that the FIFO 54 contains, instead of |S_(n−k,i)−{circumflex over (B)}_(n−k,i)| for each of the bands i, $\max\limits_{f \in {\lbrack{{f{({i - 1})}},{{f{(i)}}\lbrack}}}}{{{S_{{n - k},f} - {\hat{B}}_{{n - k},i}}}.}$

Because of the independent estimates of the long-term fluctuations {circumflex over (B)}_(n,i) and short-term variability ΔB_(n,i) ^(max) of the noise, the overestimator {circumflex over (B)}′_(n,i) makes the noise suppression process highly robust to musical noise.

The module 55 shown in FIG. 1 performs a first spectral subtraction phase. This phase supplies, with the resolution of the bands i (1≦i≦I), the frequency response H_(n,i) ¹ of a first noise suppression filter, as a function of the components S_(n,i) and {circumflex over (B)}_(n,i) and the overestimation coefficients α′_(n,i). This computation can be performed for each band i using the equation: $\begin{matrix} {H_{n,i}^{1} = \frac{\max \left\{ {{S_{n,i} - {\alpha_{n,i}^{\prime} \cdot {\hat{B}}_{n,i}}},{\beta_{i}^{1} \cdot {\hat{B}}_{n,i}}} \right\}}{S_{{n - {\tau 4}},i}}} & (7) \end{matrix}$

in which τ4 is an integer delay such that τ4≧0 (for example τ4=0). The coefficient β_(i) ¹ in equation (7), like the coefficient βp_(i) in equation (3), represents a floor used conventionally to avoid negative values or excessively low values of the noise-suppressed signal.

In a manner known in the art (see EP-A-0 534 837), the overestimation coefficient α′_(n,i) in equation (7) could be replaced by another coefficient equal to a function of α′_(n,i) and an estimate of the signal-to-noise ratio (for example S_(n,i)/{circumflex over (B)}_(n,i)), this function being a decreasing function of the estimated value of the signal-to-noise ratio. This function is then equal to α′_(n,i) for the lowest values of the signal-to-noise ratio. If the signal is very noisy, there is clearly no utility in reducing the overestimation factor. This function advantageously decreases toward zero for the highest values of the signal/noise ratio. This protects the highest energy areas of the spectrum, in which the speech signal is the most meaningful, the quantity subtracted from the signal then tending toward zero.

This strategy can be refined by applying it selectively to the harmonics of the pitch frequency of the speech signal if the latter features vocal activity.

Accordingly, in the embodiment shown in FIG. 1, a second noise suppression phase is performed by a harmonic protection module 56. This module computes, with the resolution of the Fourier transform, the frequency response H_(n,f) ² of a second noise suppression filter as a function of the parameters H_(n,i) ¹, α′_(n,i), {circumflex over (B)}_(n,i), δ_(n), S_(n,i) and the pitch frequency f_(p)=F_(e)/T_(p) computed outside silence phases by a harmonic analysis module 57. In a silence phase (δ_(n)=0), the module 56 is not in service, i.e. H_(n,f) ²=H_(n,i) ¹ for each frequency f of a band i. The module 57 can use any prior art method to analyse the speech signal of the frame to determine the pitch period T_(p), expressed as an integer or fractional number of samples, for example a linear prediction method.

The protection afforded by the module 56 can consist in effecting, for each frequency f belonging to a band i: $\left\{ \begin{matrix} {H_{n,f}^{2} = 1} & {{if}\quad \left\{ \begin{matrix} {{S_{n,i} - {\alpha_{n,i}^{\prime} \cdot {\hat{B}}_{n,i}}} > {\beta_{i}^{2} \cdot {\hat{B}}_{n,i}}} & (8) \\ {{and}\quad {\exists{{\eta \quad {{integer}/{{f - {\eta \cdot f_{p}}}}}} \leq {{\Delta f}/2}}}} & {\quad (9)} \end{matrix} \right.} \\ {H_{n,f}^{2} = H_{n,f}^{1}} & {otherwise} \end{matrix} \right.$

Δf=F_(e)/N represents the spectral resolution of the Fourier transform. If H_(n,f) ²=1, the quantity subtracted from the component S_(n,f) is zero. In this computation, the floor coefficients β_(i) ² (for example β_(i) ²=β_(i) ¹) express the fact that some harmonics of the pitch frequency f_(p) can be masked by noise, so that there is no utility in protecting them.

This protection strategy is preferably applied for each of the frequencies closest to the harmonics of f_(p), i.e. for any integer η.

If δf_(p) denotes the frequency resolution with which the analysis module 57 produces the estimated pitch frequency f_(p), i.e. if the real pitch frequency is between f_(p)−δf_(p)/2 and f_(p)+δf_(p)/2, then the difference between the η-th harmonic of the real pitch frequency and its estimate η×f_(p) (condition (9)) can go up to ±η×δf_(p)/2. For high values of η, the difference can be greater than the spectral half-resolution Δf/2 of the Fourier transform. To take account of this uncertainty, and to guarantee good protection of the harmonics of the real pitch, each of the frequencies in the range [η×f_(p)−η×δf_(p)/2, η×f_(p)+η×δf_(p)/2] can be protected, i.e. condition (9) above can be replaced with:

∃ηinteger/|f−η·f_(p)|≦(η·δf_(p)+Δf)/2  (9′)

This approach (condition (9′)) is of particular benefit if the values of η can be high, especially if the process is used in a broadband system.

For each protected frequency, the corrected frequency response H_(n,f) ² f can be equal to 1, as indicated above, which in the context of spectral subtraction corresponds to the subtraction of a zero quantity, i.e. to complete protection of the frequency in question. More generally, this corrected frequency response H_(n,f) ² could be taken as equal to a value from 1 to H_(n,f) ¹ according to the required degree of protection, which corresponds to subtracting a quantity less than that which would be subtracted if the frequency in question were not protected.

The spectral components S_(n,f) ² of a noise-suppressed signal are computed by a multiplier 58:

S _(n,f) ² =H _(n,f) ² ·S _(n,f)  (10)

This signal S_(n,f) ² is supplied to a module 60 which computes a masking curve for each frame n by applying a psychoacoustic model of how the human ear perceives sound.

The masking phenomenon is a well-known principle of the operation of the human ear. If two frequencies are present simultaneously, it is possible for one of them not to be audible. It is then said to be masked.

There are various methods of computing masking curves. The method developed by J. D. Johnston can be used, for example (“Transform Coding of Audio Signals Using Perceptual Noise Criteria”, IEEE Journal on Selected Areas in Communications, Vol. 6, No. 2, February 1988). That method operates in the barks frequency scale. The masking curve is seen as the convolution of the spectrum spreading function of the basilar membrane in the bark domain with the exciter signal, which in the present application is the signal S_(n,f) ². The spectrum spreading function can be modelled in the manner shown in FIG. 7. For each bark band, the contribution of the lower and higher bands convoluted with the spreading function of the basilar membrane is computed from the equation: $\begin{matrix} {C_{n,q} = {{\sum\limits_{q^{\prime} = 0}^{q - 1}\quad \frac{S_{n,q^{\prime}}^{2}}{\left( 10^{10/10} \right)^{({q - q^{\prime}})}}} + {\sum\limits_{q^{\prime} = {q + 1}}^{Q}\quad \frac{S_{n,q^{\prime}}^{2}}{\left( 10^{25/10} \right)^{({q^{\prime} - q})}}}}} & (11) \end{matrix}$

in which the indices q and q′ designate the bark bands (0≦q,q′≦Q) and S_(n,q) ² represents the average of the components S_(n,f) ² of the noise-suppressed exciter signal for the discrete frequencies f belonging to the bark band q′.

The module 60 obtains the masking threshold M_(n,q) for each bark band q from the equation:

M _(n,q) =C _(n,q) /R _(q)  (12)

in which R_(q) depends on whether the signal is relatively more or relatively less voiced. As is well-known in the art, one possible form of R_(q) is:

10·log₁₀(R _(q))=(A+q)·χ+B·(1−χ)  (13)

with A=14.5 and B=5.5. χ designated a degree of voicing of the speech signal, varying from 0 (no voicing) to 1 (highly voiced signal). The parameter χ can be of the form known in the art: $\begin{matrix} {\chi = {\min \left\{ {\frac{SFM}{{SFM}_{\max}},1} \right\}}} & (12) \end{matrix}$

where SFM represents the ratio in decibels between the arithmetic mean and the geometric mean of the energy of the bark bands and SFM_(max)=−60 dB.

The noise suppression system further includes a module 62 which corrects the frequency response of the noise suppression filter as a function of the masking curve M_(n,q) computed by the module 60 and the overestimates {circumflex over (B)}′_(n,i) computed by the module 45. The module 62 decides which noise suppression level must really be achieved.

By comparing the envelope of the noise overestimate with the envelope formed by the masking thresholds M_(n,q), a decision is taken to suppress noise in the signal only to the extent that the overestimate {circumflex over (B)}′_(n,i) is above the masking curve. This avoids unnecessary suppression of noise masked by speech.

The new response H_(n,f) ³, for a frequency f belonging to the band i defined by the module 12 and the bark band q, thus depends on the relative difference between the overestimate {circumflex over (B)}′_(n,i) of the corresponding spectral component of the noise and the masking curve M_(n,q) in the following manner: $\begin{matrix} {H_{n,f}^{3} = {1 - {{\left( {1 - H_{n,f}^{2}} \right) \cdot \max}\left\{ {\frac{{\hat{B}}_{n,i}^{\prime} - M_{n,q}}{{\hat{B}}_{n,i}^{\prime}},0} \right\}}}} & (14) \end{matrix}$

In other words, the quantity subtracted from a spectral component S_(n,f), in the spectral subtraction process having the frequency response H_(n,f) ³ is substantially equal to whichever is the lower of the quantity subtracted from this spectral component in the spectral subtraction process having the frequency response H_(n,f) ² and the fraction of the overestimate {circumflex over (B)}′_(n,i) of the corresponding spectral component of the noise which possibly exceeds the masking curve M_(n,q).

FIG. 8 illustrates the principle of the correction applied by the module 62. It shows in schematic form an example of a masking curve M_(n,q) computed on the basis of the spectral components S_(n,f) ² of the noise-suppressed signal as well as the overestimate {circumflex over (B)}′_(n,i) of the noise spectrum. The quantity finally subtracted from the components S_(n,f) is that shown by the shaded areas, i.e. it is limited to the fraction of the overestimate {circumflex over (B)}′_(n,i) of the spectral components of the noise which is above the masking curve.

The subtraction is effected by multiplying the frequency response H_(n,f) ³ of the noise suppression filter by the spectral components S_(n,f) of the speech signal (multiplier 64). The module 65 then reconstructs the noise-suppressed signal in the time domain by applying the inverse fast Fourier transform (IFFT) to the samples of frequency S_(n,f) ³ delivered by the multiplier 64. For each frame, only the first N/2=128 samples of the signal produced by the module 65 are delivered as the final noise-suppressed signal s³, after overlap-add reconstruction with the N/2=128 last samples of the preceding frame (module 66).

FIG. 9 shows a preferred embodiment of a noise suppression system using the invention. The system includes a number of components similar to corresponding components of the system shown in FIG. 1, for which the same reference numbers are used. Accordingly, the modules 10, 11, 12, 15, 16, 45 and 55 supply in particular the quantities S_(n,i), {circumflex over (B)}_(n,i), α′_(n,i), {circumflex over (B)}′_(n,i) and H_(n,f) ¹ used for selective noise suppression.

The frequency resolution of the fast Fourier transform 11 constitutes a limitation of the system shown in FIG. 1. The frequency protected by the module 56 is not necessarily the precise pitch frequency f_(p), but the frequency closest to it in the discrete spectrum. In some cases, harmonics relatively far away from the pitch harmonics may be protected. The system shown in FIG. 9 alleviates this drawback by appropriately conditioning the speech signal.

This conditioning modifies the sampling frequency of the signal so that the period 1/f_(p) exactly covers an integer number of sample times of the conditioned signal.

Many methods of harmonic analysis which can be used by the module 57 are capable of supplying a fractional value of the delay T_(p), expressed as a number of samples at the initial sampling frequency F_(e). A new sampling frequency f_(e) is then chosen which is equal to an integer multiple of the estimated pitch frequency, i.e. f_(e)=p·f_(p)=p·F_(e)/T_(p)=K·F_(e), where p is an integer. To avoid losing signal samples, f_(e) must be higher than F_(e). In particular, to facilitate conditioning it is possible to impose the condition that f_(e) must lie in the range from F_(e) to 2F_(e) (1≦K≦2).

Of course, it is not necessary to condition the signal if no vocal activity is detected in the current frame (δ_(n)≠0) or if the delay T_(p) estimated by the module 57 is an integer delay.

For each pitch harmonic to correspond to an integer number of samples of the conditioned signal, the integer p must be a factor of the size N of the signal window produced by the module 10: N=αp, where α is an integer. This size N is usually a power of 2 for the implementation of the FFT. It is 256 in the example considered here.

The spectral resolution Δf of the discrete Fourier transform of the conditioned signal is given by the equation Δf=p·f_(p)/N=f_(p)/α. It is therefore beneficial to make p small, to maximise α, but large enough to perform oversampling. In the example considered here, where F_(e)=8 kHz and N=256, the values chosen for the parameters p and α are indicated in table I.

TABLE I  500 Hz < f_(p) < 1000 Hz  8 < T_(p) < 16 p = 16 α = 16 250 Hz < f_(p) < 500 Hz 16 < T_(p) < 32 p = 32 α = 8  125 Hz < f_(p) < 250 Hz 32 < T_(p) < 64 p = 64 α = 4  62.5 Hz < f_(p) < 125 Hz   64 < T_(p) < 128  p = 128 α = 2  31,25 Hz < f_(p) < 62,5 Hz  128 < T_(p) < 256  p = 256 α = 1 

The choice is made by a module 70 according to the value of the delay T_(p) supplied by the harmonic analysis module 57. The module 70 supplies the ratio K between the sampling frequencies to three frequency changer modules 71, 72, 73.

The module 71 transforms the values S_(n,i), {circumflex over (B)}_(n,i), α′_(n,i), {circumflex over (B)}′_(n,i) and H_(n,f) ¹ relating to the bands i defined by the module 12 into the modified frequency scale (sampling frequency f_(e)). This transformation merely expands the bands i by the factor K. The transformed values are supplied to the harmonic protection module 56.

The latter module then operates as before to supply the frequency response H_(n,f) ² of the noise suppression filter. This response H_(n,f) ² is obtained in the same manner as in FIG. 1 (conditions (8) and (9)), except that, in condition (9), the pitch frequency f_(p)=f_(e)/p is defined according to the value of the integer delay p supplied by the module 70, the module 70 also supplying the frequency resolution Δf.

The module 72 oversamples the frame of N samples supplied by the windowing module 10. Oversampling by a rational factor K (K=K1/K2) consists in first oversampling by the integer factor K1 and then undersampling by the integer factor K2. This oversampling and undersampling by integer factors can be effected in the conventional way by means of banks of polyphase filters.

The conditioned signal frame s′ supplied by the module 72 includes KN samples at the frequency f_(e). The samples are sent to a module 75 which computes their Fourier transform. The transformation can be effected on the basis of two blocks of N=256 samples: one constituted by the first N samples of the frame of length KN of the conditioned signal s′ and the other of the last N samples of that frame. The two blocks therefore have an overlap of (2−K)×100%. For each of the two blocks, a set of Fourier components S_(n,f) is obtained. The components S_(n,f) are supplied to the multiplier 58, which multiplies them by the spectral response H_(n,f) ² to deliver the spectral components S_(n,f) ² of the first noise-suppressed signal.

The components S_(n,f) ² are sent to the module 60 which computes the masking curves in the manner previously indicated.

When computing the masking curves, the magnitude χ designating the degree of voicing of the speech signal (equation (13)) is preferably taken in the form χ=1−H, where H is an entropy of the autocorrelation of the spectral components S_(n,f) ² of the noise-suppressed conditioned signal. The autocorrelations A(k) are computed by a module 76, for example using the equation: $\begin{matrix} {{A(k)} = \frac{\sum\limits_{f = 0}^{{N/2} - 1}\quad {S_{n,f}^{2} \cdot S_{n,{f + k}}^{2}}}{\sum\limits_{f = 0}^{{N/2} - 1}{\sum\limits_{f^{\prime} = 0}^{{N/2} - 1}{S_{n,f}^{2} \cdot S_{n,{f + f^{\prime}}}^{2}}}}} & (15) \end{matrix}$

A module 77 then computes the normalised entropy H and supplies it to the module 60 for computing the masking curve (see S. A. McClellan et al.: “Spectral Entropy: an Alternative Indicator for Rate Allocation?”, Proc. ICASSP'94, pages 201-204): $\begin{matrix} {H = \frac{\sum\limits_{k = 0}^{{N/2} - 1}{{A(k)} \cdot {\log \left\lbrack {A(k)} \right\rbrack}}}{\log \left( {N/2} \right)}} & (16) \end{matrix}$

Because of the conditioning of the signal, and its noise suppression by the filter H_(n,f) ² the normalised entropy H constitutes a measurement of voicing that is very robust to noise and to pitch variations.

The correction module 62 operates in the same manner as that of the system shown in FIG. 1, allowing for the overestimated noise {circumflex over (B)}′_(n,i) rescaled by the frequency changer module 71. It supplies the frequency response H_(n,f) ³ of the final noise suppression filter, which is multiplied by the spectral components S_(n,f) of the conditioned signal by the multiplier 64. The resulting components S_(n,f) ³ are processed back to the time domain by the IFFT module 65. A module 80 at the output of the IFFT module 65 combines, for each frame, the two signal blocks resulting from the processing of the two overlapping blocks supplied by the FFT 75. This combination can consist of a Hamming weighted sum of the samples to form a noise-suppressed conditioned signal frame of KN samples.

The module 73 changes the sampling frequency of the noise-suppressed conditioned signal supplied by the module 80. The sampling frequency is returned to F_(e)=f_(e)/K by operations which are the inverse of those effected by the module 75. The module 73 delivers N=256 samples per frame. After overlap-add reconstruction using the last N/2=128 samples of the preceding frame, only the first N/2=128 samples of the current frame are finally retained to form the final noise-suppressed signal s³ (module 66).

In a preferred embodiment, a module 82 manages the windows formed by the module 10 and saved by the module 66, to retain a number M of samples equal to an integer multiple of T_(p)=F_(e)/f_(p). This avoids problems of phase discontinuity between frames. In a corresponding manner, the management module 82 controls the windowing module 10 so that the overlap between the current frame and the next corresponds to N-M. This overlap of N-M samples is taken into account in the overlap-add operation effected by the module 66 when processing the next frame. From the value of T_(p) supplied by the harmonic analysis module 57, the module 82 computes the number of samples to be retained M=T_(p)×E[N/(2T_(p))], E[ ] designating the integer part, and controls the modules 10 and 66 accordingly.

In the embodiment just described, the pitch frequency is estimated as an average over the frame. The pitch can vary slightly over this duration. It is possible to allow for these variations in the context of the present invention by conditioning the signal to obtain a constant pitch in the frame by artificial means.

This requires the harmonic analysis module 57 to supply the time intervals between consecutive breaks of the speech signal which can be attributed to glottal closures of the speaker occurring during the duration of the frame. Methods which can be used to detect such micro-breaks are well-known in the art of harmonic analysis of speech signals. In this connection, reference may be had to the following articles: M. BASSEVILLE et al., “Sequential detection of abrupt changes in spectral characteristics of digital signals”, IEEE Trans. on Information Theory, 1983, Vol. IT-29, No.5, pages 708-723; R. ANDRE-OBRECHT, “A new statistical approach for the automatic segmentation of continuous speech signals”, IEEE Trans. on Acous., Speech and Sig. Proc., Vol. 36, No.1, January 1988; and C. MURGIA et al., “An algorithm for the estimation of glottal closure instants using the sequential detection of abrupt changes in speech signals”, Signal Processing VII, 1994, pages 1685-1688.

The principle of the above methods is to effect a statistical test between a short-term model and a long-term model. Both models are adaptive linear prediction models. The value of the statistical test w_(m) is the cumulative sum of the a posteriori likelihood ratio of two distributions, corrected by the Kullback divergence. For a distribution of residues having a Gaussian statistic, the value w_(m) is given by: $\begin{matrix} {w_{m} = {\frac{1}{2}\left\lbrack {\frac{2 \cdot e_{m}^{0} \cdot e_{m}^{1}}{\sigma_{1}^{2}} - {\left( {1 + \frac{\sigma_{0}^{2}}{\sigma_{1}^{2}}} \right) \cdot \frac{\left( e_{m}^{0} \right)^{2}}{\sigma_{0}^{2}}} + \left( {1 - \frac{\sigma_{0}^{2}}{\sigma_{1}^{2}}} \right)} \right\rbrack}} & (17) \end{matrix}$

where e_(m) ⁰ and σ₀ ² represent the residue computed at the time of sample m of the frame and the variance of the long-term model, e_(m) ¹ and σ₁ ² likewise representing the residue and the variance of the short-term model. The closer the two models, the closer the statistical test value w_(m) to 0. In contrast, if the two models are far away from each other, the value w_(m) becomes negative, which denotes a break R in the signal.

Thus FIG. 10 shows one possible example of the evolution of the value w_(m), showing the breaks R in the speech signal. The time intervals t_(r) (r=1,2, etc.) between two consecutive breaks R are computed and expressed as a number of samples of the speech signal. Each interval t_(r) is inversely proportional to the pitch frequency f_(p) which is thus estimated locally: f_(p)=F_(e)/t_(r) over the r-th interval.

The time variations of the pitch (i.e. the fact that the intervals t_(r) are not all equal over a given frame), can then be corrected to obtain a constant pitch frequency in each of the analysis frames. This correction is effected by modifying the sampling frequency over each interval t_(r) to obtain constant intervals between two glottal closures after oversampling. Thus the duration between two breaks is modified by oversampling with a variable ratio, so as to lock onto the greatest interval. Also, the conditioning constraint, whereby the oversampling frequency is a multiple of the estimated pitch frequency, is complied with.

FIG. 11 shows the means employed to perform the conditioning of the signal in the latter case. The harmonic analysis module 57 uses the above analysis method and supplies the intervals t_(r) relating to the signal frame produced by the module 10. For each of these intervals, the module 70 (block 90 in FIG. 11) computes the oversampling ratio K_(r)=p_(r)/t_(r), where the integer p_(r) is given by the third column of table I if t_(r) takes the values indicated in the second column. These oversampling ratios K_(r) are supplied to the frequency changer modules 72 and 73 so that the interpolations are effected with the sampling ratio K_(r) over the corresponding time interval t_(r).

The greatest time interval T_(p) of the time intervals t_(r) supplied by the module 57 for a frame is selected by the module 70 (block 91 in FIG. 11) to obtain a pair p,α as indicated in table I. The modified sampling frequency is then f_(e)=p·F_(e)/T_(p) as previously, the spectral resolution Δf of the discrete Fourier transform of the conditioned signal still being given by Δf=F_(e)/(α·T_(p)) For the frequency changer module 71, the oversampling ratio K is given by K=p/T_(p) (block 92). The module 56 for protecting the pitch harmonics operates in the same manner as before, using for condition (9) the spectral resolution Δf supplied by the block 91 and the pitch frequency f_(p)=f_(e)/p defined according to the value of the integer delay p supplied by the block 91.

This embodiment of the invention also implies adaptation of the window management module 82. The number M of samples of the noise-suppressed signal to be retained over the current frame here corresponds to an integer number of consecutive time intervals t_(r) between two glottal closures (see FIG. 10). This avoids the problems of phase discontinuity between frames, whilst allowing for possible variations of the time intervals t_(r) over a frame. 

What is claimed is:
 1. Method of conditioning a digital speech signal processed by successive frames, comprising a harmonic analysis of the speech signal to estimate a pitch frequency of the speech signal over each frame in which the speech signal features vocal activity, and, after estimating the pitch frequency of the speech signal over one frame, conditioning the speech signal of said one frame by oversampling the speech signal in the time domain at an oversampling frequency which is an integer multiple of the estimated pitch frequency.
 2. Method according to claim 1, wherein spectral components of the speech signal are computed by distributing the conditioned signal into blocks of N samples transformed into the frequency domain, N being a predetermined integer, and wherein the ratio between the oversampling frequency and the estimated pitch frequency is a factor of the number N.
 3. Method according to claim 2, wherein the number N is a power of
 2. 4. Method according to claim 2, wherein a degree of voicing of the speech signal is estimated over the frame from an entropy of an autocorrelation of spectral components computed on the basis of the conditioned signal.
 5. Method according to claim 4, wherein the degree of voicing is measured on the basis of a normalised entropy H of the form: $H = \frac{\sum\limits_{k = 0}^{{N/2} - 1}{{A(k)} \cdot {\log \left\lbrack {A(k)} \right\rbrack}}}{\log \left( {N/2} \right)}$

where A(k) is the normalised autocorrelation defined by: ${A(k)} = \frac{\sum\limits_{f = 0}^{{N/2} - 1}\quad {S_{n,f}^{2} \cdot S_{n,{f + k}}^{2}}}{\sum\limits_{f = 0}^{{N/2} - 1}{\sum\limits_{f^{\prime} = 0}^{{N/2} - 1}{S_{n,f}^{2} \cdot S_{n,{f + f^{\prime}}}^{2}}}}$

S_(n,f) ² designating said spectral component of rank f computed on the basis of the oversampled signal.
 6. Method according to claim 1, wherein, after processing each conditioned signal frame, a number of signal samples supplied by such processing is retained which is equal to an integer multiple of the ratio between an initial sampling frequency and the estimated pitch frequency.
 7. Method according to claim 1, wherein the estimation of the pitch frequency of the speech signal over a frame includes the steps of: estimating time intervals between two consecutive breaks of the signal which can be attributed to glottal closures of speaker occurring during the frame, the estimated pitch frequency being inversely proportional to said time intervals; interpolating the speech signal in said time intervals, so that the conditioned signal resulting from such interpolation has a constant time interval between two consecutive breaks.
 8. Method according to claim 7, wherein, after processing each frame, a number of samples of the speech signal supplied by such processing is retained which corresponds to an integer number of estimated time intervals.
 9. Device for conditioning a digital speech signal processed by successive frames, comprising harmonic analysis means to estimate a pitch frequency of the speech signal over each frame in which the speech signal features vocal activity, and conditioning means for conditioning the speech signal of said frame by oversampling the speech signal in the time domain at an oversampling frequency which is an integer multiple of the estimated pitch frequency.
 10. Device according to claim 9, distributing the conditioned signal into blocks of N samples, N being a predetermined integer, and means for computing spectral components of the speech signal by transforming said blocks into the frequency domain, and wherein the ratio between the oversampling frequency and the estimated pitch frequency is a factor of the number N.
 11. Device according to claim 10, wherein the number N is a power of
 2. 12. Device according to claim 10, further comprising means for estimating a degree of voicing of the speech signal over each frame from an entropy of an autocorrelation of spectral components computed on the basis of the conditioned signal.
 13. Device according to claim 12, wherein the degree of voicing is measured on the basis of a normalised entropy H of the form: $H = \frac{\sum\limits_{k = 0}^{{N/2} - 1}{{A(k)} \cdot {\log \left\lbrack {A(k)} \right\rbrack}}}{\log \left( {N/2} \right)}$

where A(k) is the normalised autocorrelation defined by: ${A(k)} = \frac{\sum\limits_{f = 0}^{{N/2} - 1}\quad {S_{n,f}^{2} \cdot S_{n,{f + k}}^{2}}}{\sum\limits_{f = 0}^{{N/2} - 1}{\sum\limits_{f^{\prime} = 0}^{{N/2} - 1}{S_{n,f}^{2} \cdot S_{n,{f + f^{\prime}}}^{2}}}}$

S_(n,f) ² designating said spectral component of rank f computed on the basis of the oversampled signal.
 14. Device according to claim 9, wherein, after processing each conditioned signal frame, a number of signal samples supplied by such processing is retained which is equal to an integer multiple of the ratio between an initial sampling frequency and the estimated pitch frequency.
 15. Device according to claim 9, wherein the harmonic analysis means include: means for estimating time intervals between two consecutive breaks of the signal which can be attributed to glottal closures of a speaker occurring during a frame, the estimated pitch frequency being inversely proportional to said time intervals; means for interpolating the speech signal in said time intervals, so that the conditioned signal resulting from such interpolation has a constant time interval between two consecutive breaks.
 16. Device according to claim 15, wherein, after processing each frame, a number of samples of the speech signal supplied by such processing is retained which corresponds to an integer number of estimated time intervals. 